Gain control plays an important role in analog signal processing (ASP). Gain control can be achieved either by placing a variable attenuator in front of an operational amplifier (op-amp) or by changing the resistance of its feedback network. For higher signal to noise ratios (SNR) it is preferable to control the gain by varying the attenuation of the feedback network of the operational amplifier rather than to place a variable attenuator in front of it. In the first case, the noise employed in the calculation of the SNR entering the input of the operational amplifier to which a signal is applied, is entirely due to the operational amplifier itself including its feedback network. In the second case, the signal which reaches the operational amplifier input via the variable attenuator preceding the operational amplifier is attenuated by the variable attenuator, whereas the noise entering the input of the operational amplifier is enhanced by the noise contribution of the preceding variable attenuator. These two effects reduce the SNR in the second case, particularly if the attenuation is high.
High precision commercially manufactured gain controlled linear operational amplifiers, such as operational amplifier AD600/AD602 manufactured by Analog Devices, employ an attenuator in front of them. More recently, a method of employing MOS-FETs in the design of a current attenuator and varying the value of its attenuation by electronic switching between the MOS-FETs has been described in the paper "An Inherently Linear and Compact MOST-Only Current Division Technique" by K. Bult and G. Geelen, IEEE Journal of Solid State Circuits, 27 No. 123, Dec. 1992, pp. 1730-1735. The attenuator employed is of the R-2R kind, described in the book "Analog Signal Processing and Instrumentation", by Arieh Arbel, Cambridge University Press, 1980, on p. 208. FIGS. 8 and 9 of the Bult et. al. paper show a coarse and a fine section of the attenuator, respectively.
FIG. 7 of this paper shows how coarse and fine sections can be combined, and FIG. 10 of this paper shows the input and output circuits of the attenuator. This particular embodiment suffers from a number of inherent drawbacks. First, the operation of different transistors at different current levels introduces distortion into the output signal, resulting from the dependence of the dynamic resistance of each transistor on the input current signal. Second, the whole attenuator, shown in FIG. 10 of the above referenced paper, is divided into three major parts: a linear V-I converter forming the input section, which consists of a voltage amplifier such as NE5534 manufactured by NEC and the input branch of the attenuator, a current divider rendering the switched attenuator proper and a linear I-V converter forming the output stage. The drawbacks of this structure are the need for two separate operational amplifiers, the additional noise added to the signal by the linear I-V converter at the input, and last but not least, the deterioration of the SNR relating to the input of the linear I-V converter, caused by the attenuator employed in front of the linear I-V converter, as explained above.
A common method of controlling the gain of an operational amplifier employs passive linear resistors in the feedback network of the operational amplifier. The operational amplifier gain is controlled by varying the ratio of these passive resistors. Gain accuracy depends ultimately upon the ability to achieve an accurate ratio between two passive resistors, and to maintain this ratio during the manufacturing process and for different pairs of resistors over the whole range of operating temperatures. The presence of passive linear resistors satisfying these requirements complicates the manufacturing process of integrated circuits (ICs). However, resistors are linear components and lead to desired linear transfer functions from operational amplifiers.
U.S. Pat. No. 4,422,051 describes a different method of controlling the operational amplifier gain, which employs a variable attenuator comprised in the feedback path of a current amplifier. Unlike the present invention, this feedback network employs a total of four transistors, whose operation is based on the translinear principle which is explained in the paper by B. Gilbert, "Translinear Circuits: a proposed classification", Electronic Letters 11, No. 1, Jan. 9, 1975, pp. 14-16. This circuit has two major disadvantages. The first disadvantage is the distortion of the output signal which results from the unavoidable spread in the size of the transistors (width and length). The second disadvantage is stability problems in the closed loop response, resulting from the dependence of the frequency response of the feedback network's transfer function on f.sub.T of the transistors, where f.sub.T is the frequency at which the current gain of a transistor equals unity. Potential instability in the closed loop response adversely affects the bandwidth of the closed loop frequency response of the amplifier.
Chapter 1 of the book "Analog Signal Processing and Instrumentation", by Arieh Arbel, Cambridge University Press, 1980 discusses both voltage-mode and current-mode operational amplifiers.
Reference is made to FIG. 1 which shows a typical voltage-mode feedback stabilized amplifier (VMFA) 1 comprising a basic voltage amplifier 4 and a feedback network, operating in the voltage mode. The feedback network of VMFA 1 consists of resistors R.sub.f 2 and R.sub.o 3 and can be considered as a voltage divider, providing a voltage attenuation, V.sub.f /V.sub.out, which equals R.sub.o /(R.sub.f +R.sub.o). If the voltage gain A.sub.v =V.sub.out /V.sub.e of voltage amplifier 4 approaches infinity, and thus if V.sub.e approaches zero, the transfer-function of VMFA 1, V.sub.out /V.sub.in, approaches the value (R.sub.o +R.sub.f)/R.sub.o, which is the inverse value of the voltage attenuation provided by the feedback network. Hence, gain accuracy depends ultimately upon the ability to achieve an accurate ratio between two linear, passive impedances, usually resistors, and to maintain this ratio during the manufacturing process and for individual pairs of resistors over the range of operating temperatures. The presence of resistors satisfying these requirements complicates the manufacturing of integrated circuits. However, resistors are linear components and lead to desired linear transfer functions from operational amplifiers.
FIG. 2 shows a recent version of a current-mode single ended feedback stabilized operational amplifier (CMFA) 6, described in the paper "Output Stage for Current-Mode Feedback Amplifiers, Theory and Application", Arieh F. Arbel and Lavy Goldminz, "Analog Integrated Circuits and Signal Processing", 2, (3) (1992), pp. 243-255. By "single ended" a single input and a single output is meant. This distinguishes a single ended amplifier from a differential amplifier having two inputs and two outputs of opposite phase.
CMFA 6 comprises a basic current amplifier, consisting of a transimpedance amplifier 14 (an operational amplifier) having an impedance ZT, which drives a current mode output stage (COS) 10. An output terminal 15 of the basic current amplifier is connected to a feedback network 2,3.
COS 10 has a pair of voltage input terminals Y1, Y2 and a pair of current output terminals, X2, Z2. Voltage terminal Y2 is driven by an output voltage V.sub.out of output voltage terminal 24 of amplifier 14, and voltage terminal Y1 is connected to ground. The two current terminals, X2 and Z2, are interconnected by COS 10. Current terminal X2 of COS 10 (node 15) is connected to the feedback network, and current terminal Z2 is connected to a grounded load 23 having an impedance ZL. COS 10 feeds an output current I.sub.out 20 into load 23.
The transfer function of COS 10 is I.sub.out =g.sub.m V.sub.out, where I.sub.out 19 equals I.sub.out 20, V.sub.out is the output voltage of amplifier 14 at output voltage terminal 24, which is an input voltage to COS 10 applied via voltage terminal Y2, and g.sub.m is the transconductance of COS 10. V.sub.X2 is the voltage across the feedback network, as measured, for example, at node 15.
CMFA 6 can employ either an FCS (floating current source) or a CCII- (class II current conveyor) as COS 10. Both the FCS and the CCII- are building blocks in analog current mode circuit design, which are described in the above referenced paper by Arbel and Goldminz.
Both the FCS and the CCII-, which can be employed as COS 10 as described above, are characterized by the fact, that the feedback current entering the first current terminal equals the output current leaving the second current terminal. This means that feedback current I.sub.out 19 entering current terminal X2 equals output current I.sub.out 20 flowing out of current terminal Z2. Hence the term "floating". The CCII- has additional features: whereas the voltage at terminal X2 is arbitrary for an FCS, it equals the voltage at terminal Y2 for an ideal CCII-. Furthermore, the output impedance at node X2 is high in the case of an FCS but low in the case of a CCII-. The choice of employing either an FCS or a CCII- as COS 10 has no bearing on the basic operation of the feedback network but both are mentioned here, since it will be shown that both the FCS and the CCII- have to be modified in order to be employed as COS 10 in preferred embodiments of the invention.
The open loop current gain of the basic current amplifier, A.sub.i =I.sub.out /I.sub.e, equals ZTg.sub.m, where ZT is the transimpedance of amplifier 14 and g.sub.m is the transconductance of COS 10, as defined above. Since inverting input terminal X1 of basic current amplifier 14 is ideally at zero potential both DC wise and for an AC signal, the feedback network of CMFA 6 can be considered as a current divider, having attenuation I.sub.f /I.sub.out which equals R.sub.o /(R.sub.f +R.sub.o), where I.sub.out equals I.sub.out 19 and I.sub.out 20. If A.sub.i of operational amplifier 14 approaches infinity, I.sub.e approaches zero and the closed loop transfer function of CMFA 6 approaches the inverse value of the feedback attenuation, i.e. I.sub.out /I.sub.in approaches (R.sub.f +R.sub.o)/R.sub.o. Resistor R.sub.f 2 will be identified herein as the feedback resistor and resistor R.sub.o 3 as the gain setting resistor.
Hence, the current transfer function of CMFA 6 comprising a basic current amplifier having a substantially infinite current gain A.sub.i, equals (R.sub.o +R.sub.f)/R.sub.o. R.sub.f 2 and R.sub.o 3 are driven in parallel both with respect to a DC and an AC signal.
In all following figures, as in FIG. 2, V.sub.X2 represents the voltage applied by the basic current amplifier across the feedback network, as measured, for example, at junction 15, and I.sub.e represents the error current which acts as an input signal to amplifier 14. I.sub.f represents the feedback current flowing through resistor R.sub.f 2 or through the non-linear element or elements substituting R.sub.f 2, comprised in preferred embodiments of the invention. I.sub.out 19 represents the total current flowing out of the feedback network and entering current terminal X2 of COS 10. I.sub.out 19 entering current terminal X2, which equals output current I.sub.out 20, flowing out of current terminal Z2 of COS 10 into load 23, equals -V.sub.X2 /(R.sub.o .vertline..vertline.R.sub.f) which equals -V.sub.X2 (R.sub.o +R.sub.f)/R.sub.o R.sub.f, where R.sub.o .vertline..vertline.R.sub.f is the equivalent resistance of R.sub.o and R.sub.f connected in parallel.